Operational amplifier

An operational amplifier is a differential electronic amplifier whose unloaded voltage is modeled in its linear regime by $v_o=A(s)(v_{+}-v_{-})$. Its large open-loop gain, large input impedance and small output impedance make negative feedback useful for setting a reproducible transfer function with external components. Supply voltages bound the available output swing; the linear model does not apply after saturation.

For an ideal operational amplifier with grounded non-inverting input, connect an input capacitor $C$ to the inverting input and a feedback resistor $R$ from output to that input. The virtual ground and Kirchhoff's current law give $C{\dot{V}}_{\mathrm{in}}=-V_{\mathrm{out}}/R$. The ideal response amplifies high-frequency noise, so practical differentiators restrict its operating band and maintain feedback stability.

An inverting operational amplifier can combine currents from several input resistors at one virtual ground. Kirchhoff's current law and negligible amplifier input current give the displayed weighted sum. Replacing the feedback resistance by a capacitor gives a weighted op amp integrator. The formulas assume stable negative feedback and operation within the amplifier's linear output range.

The closed-loop gain from an equivalent differential input-voltage disturbance to the output is the noise gain. For an ideal-input inverting amplifier, $\beta =R_1/(R_1+R_2)$ and $G_{\mathrm{n}}=1+R_2/R_1$, although the voltage gain is $-R_2/R_1$. A dominant-pole amplifier has approximate closed-loop bandwidth $f_T/G_{\mathrm{n}}$ when loop gain is large at low frequency and the circuit is stable.

A non-inverting operational amplifier applies the signal to its non-inverting input and returns part of the output to its inverting input through negative feedback. Connect resistor $R_f$ from the output to the inverting input, and resistor $R_g$ from that input to a reference circuit node at electric potential $V_{\mathrm{ref}}$.

In stable, unsaturated ideal operation, the inputs draw no electric current and their electric potentials are equal. Writing the signal as $V_{\mathrm{in}}$, Kirchhoff's current law at the inverting input givessoFor a grounded reference circuit node, the voltage gain is $1+R_f/R_g$. For example, $R_f=20\mathrm{k\Omega }$ and $R_g=10\mathrm{k\Omega }$ give gain three. The equal-input-potential approximation requires sufficiently large loop gain; amplifier saturation or unstable feedback invalidates it.^[ref]

In amplifier saturation, the requested output lies outside the amplifier's linear operating range, such as when supply rails limit output voltage. The negative feedback virtual short approximation then need not hold. Output swing, current drive and recovery behaviour depend on the device and load; finite slew rate is a separate dynamic limit even when the output has not reached a rail.

Return a fraction of the output to the inverting input so that it opposes the input difference. Large stable loop gain makes the closed-loop gain depend mainly on the external feedback ratio rather than the uncertain open-loop gain. Frequency-dependent phase lag can destabilize the loop; saturation and finite slew rate also limit the approximation.^[ref]

The closed-loop gain is the input-to-output transfer after closing the feedback connection. In the elementary noninverting voltage model it tends to $1/\beta$ for large stable loop gain, rather than retaining the uncertain open-loop gain $A$. A frequency-dependent transfer must be checked for stability before that approximation is used.

The loop gain is the product of the forward amplifier response and the return-network fraction, with the summing-junction minus sign kept explicit. Closed-loop poles are determined by $1+L=0$ in this convention. Large low-frequency magnitude controls gain error, while frequency-dependent phase affects stability.

The phase margin measures the remaining phase lag before a negative-feedback loop reaches the critical phase $-\pi$ at unity loop magnitude, using a continuous phase convention. Positive adequate margin is a useful stability and transient-response design diagnostic for standard single-crossover amplifier loops. More general unstable open-loop systems and multiple crossovers require the full stability criterion.

A voltage-feedback operational amplifier driven at its noninverting input, with a feedback divider $R_f,R_g$ at its inverting input, has ideal gain $1+R_f/R_g$ under stable negative feedback.

A voltage-feedback operational amplifier with its noninverting input grounded and input and feedback resistances $R_i,R_f$ has ideal gain $-R_f/R_i$. Virtual short and negligible input current give the result by node-current balance.

An ideal operational amplifier with a series input resistor and capacitor, resistive feedback and a grounded noninverting terminal realizes the displayed high-pass filter. Virtual ground and Kirchhoff's current law give $H=-R_f/[R_i+1/(i\omega C)]$. Its phase is $-\pi /2-{\tan }^{-1}(\omega R_{i}C)$ in the $e^{i\omega t}$ convention. Equal resistances do not make its phase frequency independent.

With the noninverting input grounded, a differential input impedance $R_{\mathrm{in}}$ draws current from the summing node. For zero output impedance, $V_o=-Av$ and Kirchhoff's current law gives $(v-V_i)/R_1+(v-V_o)/R_2+v/R_{\mathrm{in}}=0$. Eliminating $v$ yields the displayed voltage gain. The resistor-ratio result requires $|A|\gg 1+R_2/R_1+R_2/R_{\mathrm{in}}$ as well as stable, unsaturated operation. The same expression applies to small-signal phasors with complex $A(f)$.

An ideal inverting operational amplifier with series $R_1,C_1$ input and parallel $R_2,C_2$ feedback has the displayed transfer function, where $K=R_2/R_1$ and ${\tau }_i=R_i{C}_i$. Its peak is at ${\omega }_0=1/\sqrt{{\tau }_1{\tau }_2}$ with voltage gain $K{\tau }_1/({\tau }_1+{\tau }_2)$. A wide input-high-pass/feedback-low-pass separation produces a broad plateau, not a narrow resonance. Validity requires stable negative feedback, sufficient loop gain and linear output operation.

For a response proportional to $s/[(1+s{\tau }_1)(1+s{\tau }_2)]$ with positive time constants, its half-power width is $1/{\tau }_1+1/{\tau }_2$. Combining with the geometric-mean centre frequency gives the displayed quality factor; the arithmetic-geometric mean inequality proves the bound. Equality requires matched time constants and gives half-power frequencies $(\sqrt{2}\pm 1){\omega }_0$. Adjusting gains does not make this topology a high-Q resonant filter.

An ideal operational amplifier with a grounded noninverting input, input resistor $R_i$, and parallel feedback resistor $R_f$ and capacitor $C$ implements a low-pass filter. The virtual short and Kirchhoff's current law give $V_{\mathrm{out}}/V_{\mathrm{in}}=-Z_f/R_i$, where $Z_f=R_f/(1+i\omega R_{f}C)$. The DC gain magnitude is $R_f/R_i$ and the cutoff frequency is $1/(2\pi R_{f}C)$. The capacitance reduces high-frequency feedback electrical impedance, causing the gain magnitude to fall.

Under stable negative feedback and within its linear operating range, an ideal voltage-feedback operational amplifier has equal input voltages. This relation is a consequence of large open-loop gain, and does not imply current flow between the inputs.

An op amp integrator places a resistor $R$ between the input source and the inverting input of an operational amplifier, and a feedback capacitor $C$ between that input and the output, with the non-inverting input grounded. For an ideal operational amplifier, Kirchhoff's current law gives $v_{\mathrm{in}}/R=-Cd{v}_o/dt$, henceThe initial capacitor charge supplies the integration constant. A parallel feedback resistor $R_f$ replaces the transfer by $H(s)=-(R_f/R)/(1+s{R}_{f}C)$: the direct-current gain becomes finite while integration is recovered when $|s{R}_{f}C|\gg 1$. This limits offset-driven saturation. ^[ref]

Consider an op amp integrator with infinite input impedance, an internal controlled voltage $u=-A(s)v$ where $v$ is the inverting-input voltage, and a series output impedance equal to a resistance $r_o$. Assume no external output load. Writing $y$ for the output, Kirchhoff's current law givesEliminating $u$ and $v$ yieldsThe expression at $r_o=0$ is obtained by its limit. In particular, constant gain $A_0$ and $r_o=0$ give $H(s)=-A_0/[1+sRC(1+A_0)]$: a finite direct-current gain followed by an integrating regime. For the dominant-pole model $A(s)=A_0/(1+s/{\omega }_a)$ and $r_o=0$,If $A_0\gg 1$ and $A_0{\omega }_{a}RC\gg 1$, there are separated turnover frequencies near $1/(A_{0}RC)$ and $A_0{\omega }_a$. The gain magnitude is respectively constant, proportional to ${\omega }^{-1}$ and proportional to ${\omega }^{-2}$. With the convention $e^{i\omega t}$ a continuous phase choice passes from $180^{\circ }$ through $90^{\circ }$ to $0^{\circ }$. If $r_o>0$ and $A(i\omega )\to 0$, capacitive feedthrough instead gives the ultimate limit $r_o/(R+r_o)$. Specifying finite gain alone therefore does not specify a unique high-frequency response.

The slew rate bounds the large-signal output-voltage change per unit time. It is distinct from a small-signal gain-bandwidth product. A sinusoidal output of amplitude $V_p$ and angular frequency $\omega$ needs $\omega V_p$ below the available slew rate to avoid this source of distortion.

An input bias current is a direct-current component drawn by an input terminal of a real operational amplifier. Through a source resistance it produces a voltage error, and through the feedback capacitor of an op amp integrator it can produce an output ramp. Its magnitude and sign depend on the amplifier technology and operating conditions.

The input offset voltage of an operational amplifier is the differential direct-current input voltage needed to make its output zero at a specified operating point. Its effect resembles a small unwanted input signal and can cause sustained output drift in an op amp integrator.

The open-loop gain of an operational amplifier is the ratio of its output voltage to its differential input voltage before an external feedback control connection is applied. A common small-signal model is $A(s)=A_0/(1+s/{\omega }_a)$, with a large direct-current gain $A_0$ and a dominant pole at $-{\omega }_a$. Above ${\omega }_a$ its magnitude falls approximately inversely with angular frequency.

For a dominant-pole voltage-feedback operational amplifier with $A(s)=A_0/(1+s/{\omega }_a)$ and $A_0\gg 1$, the unity-gain angular frequency is approximately ${\omega }_t=A_0{\omega }_a$. In the range where this pole dominates, a roughly constant closed-loop noise gain $G_N$ gives a bandwidth approximately $f_t/G_N$. This approximation depends on the amplifier model and feedback control circuit; it is not a universal statement about all amplifiers or all frequency-dependent feedback networks. ^[ref]

An ideal operational amplifier has infinite open-loop gain and input impedance, zero output impedance, unlimited response speed and zero input offset voltage and input bias current. Its input currents vanish. In a stable linear circuit with negative feedback and a finite output, its differential input voltage tends to zero as the open-loop gain tends to infinity. Thus $i_{+}=i_{-}=0$ and $v_{+}=v_{-}$ are the familiar conditional circuit rules. The equality of the input voltages cannot be imposed on an open-loop, unstable or saturated circuit.

Connect a resistor $R_{\mathrm{in}}$ from the input source to the inverting input, a feedback resistor $R_f$ from output to that same input, and ground the non-inverting input. For ideal linear operation the inverting node is a virtual ground. Kirchhoff's current law gives $V_i/R_{\mathrm{in}}+V_o/R_f=0$, yielding the displayed voltage gain. The source sees input impedance $R_{\mathrm{in}}$, even though the amplifier input itself has infinite input impedance.