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A current-plane convention in heterodyne calibration specifies which electrical current an inferred beat amplitude denotes: the detector's generated small-signal current, the current entering a cable, or the current dissipating power in a particular load. Those amplitudes need not be equal. A power-to-current conversion is meaningful only with its circuit and reference plane.

An original photodiode model uses generated peak-current phasor $I_d$, detector shunt admittance $Y_d$ and load impedance $Z_L$. Kirchhoff's current balance givesThe factor one half assumes peak phasors; with RMS phasors it disappears. The current in the load determines delivered electric power. The generated current is recovered only after correcting the current division and any intervening network.

For a resistive Norton model with source resistance and load resistance both $R$, $Y_d=1/R$, so $I_L=I_d/2$ and $P_L=|I_d{|}^{2}R/8$. If the source shunt admittance is negligible, $I_L\approx I_d$ instead. Calling both circuits “a detector feeding a 50-ohm instrument” does not establish the same current conversion.

Capacitive current division in heterodyne receivers adds frequency dependence. With $Y_d=i\omega C_d$ and resistive load $R$,For stipulated $C_d=1$ pF and $R=50\Omega$, this magnitude is approximately 0.9540 at 1 GHz and 0.3033 at 10 GHz. The same generated amplitude then gives a smaller measured load current at the higher frequency. These are equivalent-circuit calculations, not measured specifications of a particular photodiode.

NIST's primary optical-receiver experiment distinguishes microwave power delivered to a reference load from the power indicated by a sensor. It corrects sensor calibration, interconnecting-network loss and impedance mismatch using measured scattering and reflection parameters.^[ref] That apparatus makes the reference plane an experimental calibration choice rather than a label applied after analysis.

Document the detector model, load, frequency, cable and connector corrections, and peak-versus-root mean square convention before normalizing by separate arm currents. If the arm currents describe generated DC photocurrent while the beat numerator describes uncorrected load current, the resulting overlap includes electrical current division. Heterodyne amplitude-versus-power gain conversion and impedance matching address different parts of that chain and both may be needed.

Heterodyne amplitude-versus-power gain conversion translates a calibration of measured electric power into a calibration of beat-current amplitude. In a fixed resistive load, power is quadratic in current amplitude, so a multiplicative power correction becomes a square-root amplitude correction. The electrical reference plane and peak-versus-root mean square convention must also agree.

For sinusoidal load current $i(t)=B\cos \omega t$ and resistance $R$,For two measurements of the same load, $P_2/P_1=(B_2/B_1{)}^2$. A power multiplier of four therefore means an amplitude multiplier of two. In decibels, the same change is $10{\log }_{10}4=20{\log }_{10}2\approx 6.0206$ dB. NIST distinguishes the factor ten for power ratios from the factor twenty for field-amplitude ratios when their quadratic relationship applies.^[ref]

If the load resistance also changes, the more general relation isFor frequency-dependent complex impedances, use the appropriate circuit and power model rather than replacing resistance by an arbitrary complex number. A measured electric potential difference likewise requires its stated impedance before it can be converted to current.

For a corrected detector-current peak amplitude $B_d$, the normalized overlap is $\gamma =B_d/(2\sqrt{I_1{I}_2})$. If the numerator is instead detector-current RMS amplitude, the denominator is $\sqrt{2I_1{I}_2}$. Using an RMS numerator with the peak denominator lowers the result by $1/\sqrt{2}$, a 3.0103 dB discrepancy under the amplitude-ratio convention. The mistake is a convention mismatch, not evidence of worse optical overlap.

Current-plane convention in heterodyne calibration is another independent issue. In an ideal matched Norton circuit, a source peak current $B_d$ splits equally between source resistance and equal load resistance $R$. Load peak current is $B_d/2$, so delivered power is $B_d^{2}R/8$, rather than $B_d^{2}R/2$. The factor depends on the specified equivalent circuit; real detector capacitance and interconnecting networks require their measured response.

NIST's two-laser receiver apparatus measures microwave power and separately corrects sensor calibration, network loss and impedance mismatch. Its matched optical modes make the modulation calculable, while the electrical corrections locate the power at the intended reference plane.^[ref] Record raw power, correction factors, reference plane, load, amplitude convention and DC-current calibration separately. Only then apply the gain-matching condition for heterodyne normalization.

John D. Ramboz is the first author, with Oskars Petersons, of NIST's 1991 account of a laboratory current transformer calibration service. The report describes ratio and phase measurements, their uncertainties and quality-control procedures for electrical measurement work.^[ref] It is a primary technical record of the service and its measurement conventions.

The separation of magnitude and phase has a direct consequence for electric power measurements. Consider an original sinusoidal model with root-mean-square electric potential difference $V$, secondary current magnitude $I_s$, true current ratio $K$ and primary-current lag $\varphi$. True active power isIf the secondary leads the primary by $\beta$, simply multiplying the secondary signal by nominal ratio $K_n$ givesprovided $\cos \varphi \ne 0$. This uses the lead-positive current-transformer phase displacement convention stated in the report.^[ref]

With exact magnitude conversion $K_n=K$ and small $\beta$ in radians, the relative power error is approximately $\beta \tan \varphi$. For stipulated $\beta =0.001$ rad and $\varphi =60^{\circ }$, that term is about $0.173\%$. At $\varphi =0$, its first-order value vanishes. Thus the same phase error can have different effects at different power factors.

This derivation is an explanatory example, not a reported performance test or an attribution of the calculation to Ramboz. It explains why a calibration report needs operating conditions, a sign convention and measurement uncertainty, in addition to a nominal transformer ratio. Precision in the current magnitude does not remove phase-dependent power error.

Assume a passive linear electrical circuit in sinusoidal steady state, with transients absent and $V_0$ the peak voltage. Write $Z=|Z|e^{i\varphi }=R+iX$. The electric current is $I(t)=I_0\cos (\omega t-\varphi )$, with $I_0=V_0/|Z|$. Average the instantaneous electric power $V(t)I(t)$ over a full period. The double-frequency term averages to zero, leavingThereforeEquivalently, root mean square values give $⟨P⟩=V_{\mathrm{rms}}{I}_{\mathrm{rms}}\cos \varphi$. The reactance exchanges stored electric field or magnetic field energy with the source, whereas the real part produces Joule heating. A purely imaginary complex impedance dissipates no mean electric power under these ideal assumptions.

A reference resistor is a resistor whose assigned electrical resistance supplies the comparison value in a resistance-measurement circuit. Its value may come from measurement calibration rather than its nominal marking. The useful reference is its value under the stated temperature, loading and connection conditions, together with the associated measurement uncertainty.

In a shared-excitation resistance ratio measurement, the sensor and reference resistor carry the same electric current. Measuring their electric potential differences gives $r=V_T/V_{\mathrm{ref}}=R_T/R_{\mathrm{ref}}$, so $R_T=r{R}_{\mathrm{ref}}$. A reference value error therefore sets a scale error even when excitation cancels perfectly. Reference-resistor error in ratiometric resistance measurement derives the exact relation and distinguishes a correction from its remaining measurement uncertainty.^[ref]

The connections define which resistance is compared. In four-terminal sensing, two leads carry the electric current while a separate high-impedance pair senses the electric potential difference at the reference terminals. This avoids including the current-lead drops in the assigned reference value, within the sensing instrument's limitations.

For an original ideal example, a $100\Omega$ reference resistor carries $1\mathrm{mA}$ through two current leads, each of electrical resistance $0.20\Omega$. The reference element has a potential difference of $0.1000\mathrm{V}$, while a reading outside both leads is $0.1004\mathrm{V}$. If this latter reading is used as the denominator while the calculation assumes $100\Omega$, a true $50\Omega$ sensor is inferred asSensing directly at the reference terminals removes this idealized lead contribution. It does not correct a wrong reference value. The circuit guide explicitly shows a Kelvin connection at the reference terminals for this purpose.^[ref]

A small temperature change shifts the value approximately by $\Delta R_{\mathrm{ref}}/R_{\mathrm{ref}}\simeq {\alpha }_R\Delta T$, with temperature coefficient of resistance ${\alpha }_R$. Excitation also dissipates electric power $I^2{R}_{\mathrm{ref}}$; a reference with a different temperature or load from its measurement calibration may require a correction or additional measurement uncertainty. A proposed check compares the reference before and after the measurement sequence under the same connection and loading conditions. Repeating the comparison at a lower electric current tests for a heating contribution. Stability of the observed ratio alone cannot certify the reference's assigned value.

A shared-excitation resistance ratio measurement compares two electrical resistances by passing the same electric current through them and comparing their electric potential differences. It is the circuit realization of a ratiometric measurement: the unknown excitation becomes a common multiplicative input.

For a sensor $R_T$ in series with a reference resistor $R_{\mathrm{ref}}$, ideal high-input-impedance sensing givesThe electric current must be nonzero and shared at the time relevant to each sample. One implementation feeds $V_T$ to an analog-to-digital converter input and $V_{\mathrm{ref}}$ to its external reference pins; the conversion is then scaled by their ratio. The primary circuit guide shows this wiring and includes the programmed gain when translating a digital code to resistance.^[ref]

For an original check, choose $R_T=180\Omega$ and $R_{\mathrm{ref}}=1200\Omega$. Excitation at $0.4\mathrm{mA}$ gives $0.072\mathrm{V}$ and $0.480\mathrm{V}$; excitation at $1.2\mathrm{mA}$ gives $0.216\mathrm{V}$ and $1.440\mathrm{V}$. Both give $r=0.15$ and recover $180\Omega$. This calculation assumes unchanged resistances; the larger excitation dissipates nine times the sensor electric power, making sensor self-heating in a ratiometric resistance measurement an independent test.

A common supply is insufficient if different branch currents pass through the two resistances. With currents $I_T,I_{\mathrm{ref}}$,This is branch-current mismatch in a resistance ratio measurement. Sequential sampling of a changing excitation has the same problem unless the two paths retain the same effective excitation history. Ratiometric cancellation with matched signal and reference filters specifies the dynamic condition.

Wiring matters even with an ideal shared current source. A two-wire sensor reading includes lead electrical resistance; four-terminal sensing separates the current leads from high-impedance potential-sensing leads. The reference itself needs the same care if lead drops would otherwise enter its measured denominator.^[ref]

A bench check can replace the sensor with a stable known resistor, sweep excitation, and compare each measured potential with their ratio. A constant ratio accompanied by changing individual potentials tests shared excitation. It does not establish reference accuracy, eliminate sensor heating, or prove operation outside the converter's input and reference ranges.

Thermal loading by a contact temperature sensor is the disturbance of an object's temperature caused by exchanging heat with the temperature sensor used to measure it. A sensor initially colder than a small object draws heat from the object: the final reading can be accurate for the disturbed system while differing from the object's original temperature.

An illuminating idealization treats the object and sensor as two uniform bodies with constant heat capacities $C_o$ and $C_s$, joined by thermal conductance $G$ and isolated from everything else. Their energy conservation equations areAdding the equations shows that $C_o{T}_o+C_s{T}_s$ is constant. Consequently,For $C_o=0.20J/K$, $C_s=0.020J/K$, $T_{o0}=80{}^{\circ }\mathrm{C}$ and $T_{s0}=20{}^{\circ }\mathrm{C}$, the common equilibrium is $74.55{}^{\circ }\mathrm{C}$. The measurement has cooled the object by about $5.45\mathrm{K}$.

Subtracting the two differential equations gives an exponentially decaying temperature difference with time constantIncreasing the contact thermal conductance makes the disturbance happen faster without changing this isolated model's final equilibrium. Reducing the sensor heat capacity reduces the disturbance. A real object also exchanges heat with its surroundings, while sensor leads can provide another heat transfer path.

An electrically read sensor can additionally generate Joule heating. That is a separate loading mechanism: the thermometry guide measures resistance-thermometer self-heating by changing the measuring electric current and extrapolating toward zero dissipated electric power.^[ref]