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An analog switch uses a control signal to connect or disconnect an analog signal path. The control chooses the conducting state; the transmitted electric potential difference is not restricted to a binary logic value. Real devices have finite analog switch on-resistance, leakage and parasitic capacitance. The primary tutorial describes a bilateral CMOS switch formed from complementary transistors.^[ref]

For an original loading example, connect an ideal $1\mathrm{V}$ source through a stipulated $100\Omega$ on-resistance to a $10\mathrm{k\Omega }$ load. The output isThe closed switch therefore changes the amplitude even before any switching transient is considered. This is a chosen resistor model, not a specification for a particular part. Variation of on-resistance with signal level can additionally produce distortion.

An open switch is also not an ideal infinite impedance. For example, a stipulated $10\mathrm{nA}$ analog switch leakage current into $100\mathrm{pF}$ changes the node potential at $I/C=100V/s$, or $0.10\mathrm{V}$ in a millisecond if that model remains valid. A high-impedance measurement can be sensitive to this small current.

On a bench, compare the transmitted level and the open-state drift with the same source, load and oscilloscope conditions. Analog switch charge injection concerns the control transition itself, while settling time concerns how long the complete circuit needs to approach its final value. Those effects are separate from the static conducting-state error.

An observation interval of finite duration is not an infinitely sharp frequency selector. Combine the lock-in quadratures intoFor unwanted electric potential difference $s(t)=B\cos ((\omega +\delta )t+\psi )$, where $\delta$ is an angular frequency offset and $\omega +\delta >0$, expansion into complex exponentials gives the exact responseThe sinc function formula follows by integrating the exponential function and factoring $e^{\mathrm{i}\nu T}-1=2\mathrm{i}{e}^{\mathrm{i}\nu T/2}\sin (\nu T/2)$; its value at zero follows from a limit. The first term changes slowly after multiplication; the second oscillates at the sum frequency. It vanishes at exact matching when $T$ is an integer number of reference cycles, each lasting one period of a function, but cannot generally be discarded in a short-window calculation.

For $|\delta |\ll \omega$ and many reference cycles, the first term usually dominates: its modulus is $B|\mathrm{sinc}(\delta T/2)|$ and its complex argument includes $\delta T/2$. Thus $|\delta |T\ll 1$ gives almost no rejection. A longer observation separates closer frequencies, with the first zeros of this term at $|\delta |=2\pi /T$. Those zeros are not a brick-wall cutoff: the sinc function has nonzero lobes beyond them. At $\delta =0$ a synchronized contribution remains however large $T$ becomes.

For a numerical check, use $f_0=1\mathrm{Hz}$, $T=4\mathrm{s}$, $\psi =0$ and an unwanted frequency $f=1.125\mathrm{Hz}$. Then $(f-f_0)T=1/2$ and the exact expression giveswhereas the difference-frequency term alone gives $2/\pi \approx 0.636620$. An interference amplitude $B=0.9\mathrm{V}$ therefore contributes $0.539254\mathrm{V}$ to $|Z_T|$ in this constructed example. A nearby frequency can survive strongly even though the three-times-frequency component in lock-in harmonic rejection cancels exactly.

A practical lock-in amplifier's low-pass filter weights past products differently from this rectangular observation interval. Its time constant and roll-off determine near-reference rejection and settling time; narrowing detection slows the response.^[ref]

An integrator reset establishes the charge on an integrating amplifier's feedback capacitor before the next measurement. Without it, the next output includes the previous initial condition, and repeated input pulses or input offset voltage can eventually cause amplifier clipping. Resetting makes consecutive measurement intervals comparable; it does not reconstruct a pulse already lost to amplifier clipping.

Consider an inverting integrating amplifier with the input held at zero during reset. Place an analog switch across its feedback capacitor $C$. In the approximation of a stable ideal operational amplifier maintaining a virtual ground, a closed switch with constant on-state electrical resistance $R_{\mathrm{on}}$ gives the RC circuit relationThis exponential decay gives an integrator reset settling time for residual magnitude at most $\epsilon$:For stipulated values $C=10\mathrm{nF}$, $R_{\mathrm{on}}=10\mathrm{k\Omega }$ and $|V_{\mathrm{old}}|=0.50\mathrm{V}$, the reset time constant is $100\mu \mathrm{s}$. Reaching a residual of $1\mathrm{mV}$ requires about $621\mu \mathrm{s}$ in this model. Actual amplifier drive, stability and recovery from overload can change the response. A switch's control delay also differs from the full circuit's settling time.^[ref]

Opening the reset switch can itself disturb the initial condition. Analog switch charge injection transfers a small charge to the connected capacitor, producing a reset charge-injection pedestal of magnitudeFor a stipulated $10\mathrm{pC}$ deposited on $10\mathrm{nF}$, this step is $1\mathrm{mV}$. Its sign depends on the circuit and charge direction. A longer reset suppresses the preceding exponential residual but need not suppress the new step created when the switch opens. Analog Devices illustrates this capacitive coupling mechanism and the resulting output step separately from retained-charge settling.^[ref]

A practical experiment repeats zero-input measurements with several reset durations and reads the output at the same delay after switch opening. A decreasing residual indicates incomplete discharge; a persistent offset can instead include reset charge-injection pedestal, input offset voltage and subsequent drift. Repeated readings help separate the reproducible offset from noise. The ACF2101 provides a concrete historical implementation: separate hold and reset controls disconnect the input and discharge the integration capacitor, and its datasheet shows the sequence of hold, integration and reset states.^[ref]

The reset interval also creates integrator reset dead time: charge arriving while the circuit is being reset may not contribute to the next measured integral. Pulse timing relative to the reset therefore matters. A perfectly repeatable baseline alone does not establish that every input pulse was captured.

A low-pass filter preferentially transmits low frequencies and attenuates higher frequencies. It can smooth rapid noise in a slowly varying signal, but it also suppresses wanted variations outside the useful signal bandwidth. An RC low-pass filter gives a concrete continuous-time example; a moving average implements related smoothing on a discrete-time signal.

The useful question is how much change the measurement can tolerate. Preserving the low frequencies of a temperature record while reducing a fast unwanted oscillation can improve readability. Preserving the sharp onset of a pulse requires higher frequencies, so excessive smoothing changes its timing and shape. The RC low-pass frequency response and RC low-pass step response make this signal bandwidth versus settling time tradeoff quantitative.

A measurement procedure describes in sufficient detail how a measurement is performed, including its measurement method, measurement model and the calculations that produce a measurement result. The VIM notes that the description usually enables an operator to carry out the measurement and can include a target measurement uncertainty.^[ref] Naming an instrument or writing a final equation alone leaves much of this operational description unspecified.

Consider an original procedure for the resistance measuring system described above. Specify the test resistor and temperature, the two-terminal lead arrangement, source current, readout settings and settling time. Record the measured or assigned current with its measurement uncertainty. First replace the test resistor by a short whose resistance is negligible in the chosen model, and record indication $V_s$. Insert the test resistor without changing the declared lead configuration, allow the specified settling interval and record $V_x$. Repeating the short reading after the sample reading helps check stability.

Assume the current $I$, lead resistance $R_L$ and additive readout offset $V_0$ remain common to the paired readings. Thenso the procedure's calculation isFor stipulated $V_s=3\mathrm{mV}$, $V_x=103\mathrm{mV}$ and $I=1\mathrm{mA}$, this gives $100\Omega$. The paired measurement removes the common lead and zero contributions under the stated model.

The sequence matters because those contributions can change. If the readout offset rises by $0.5\mathrm{mV}$ between the short and sample readings, the same calculation is shifted by $0.5\mathrm{mV}/(1\mathrm{mA})=0.5\Omega$. This is an original instrumental drift example, not observed equipment performance.

Document which operations are repeated, how corrections are obtained, how deviations are handled and which uncertainty contributions remain. The procedure turns the apparatus and model into a repeatable set of operations; it does not make the supplied references, stability assumptions or recorded values exact.

A positioning reversal error compares the mean reached positions at one command when approached from opposite directions. Define $e_{i,j,d}=x_{i,j,d}-x_i^{\ast }$ for target $i$, trial $j$ and approach direction $d$. With $\uparrow$ denoting approach in the increasing-coordinate direction, use the signed target-specific quantityState this sign convention. PI's axis-level quantity called maximum positioning backlash instead takes the maximum absolute directional mean difference over the tested targets.^[ref]

For an original two-target example, measured mean errors at $x^{\ast }=2\mathrm{mm}$ are $0.3\mu \mathrm{m}$ from below and $-0.7\mu \mathrm{m}$ from above, giving $B_1=1.0\mu \mathrm{m}$. At $x^{\ast }=8\mathrm{mm}$, means of $-0.4$ and $0.9\mu \mathrm{m}$ give $B_2=-1.3\mu \mathrm{m}$. ThusA small signed average can conceal substantial reversal effects that change sign along the travel.

Read final load position independently, for example with an interferometer, after an agreed settling time. The observed difference does not identify a unique physical cause: backlash, elastic relaxation or instrumental drift can contribute. For an original drift model $e_d(t)=a_d+vt$, an unbalanced timing schedule givesAt $v=0.01\mu \mathrm{m}{\mathrm{s}}^{-1}$ and mean measurement times 50 s and 150 s, drift alone contributes $-1\mu \mathrm{m}$. Alternate or balance the approach timing when testing that confounding effect. Unidirectional positioning repeatability measures within-direction scatter rather than this difference of means.

Range-overlap calibration establishes or checks the relationship between instrument settings using inputs that both can measure validly. In gain ranging, it tests whether changing sensitivity preserves the inferred measurand. It can determine relative scale and offset between ranges, but agreement between ranges alone does not establish their absolute accuracy or independent metrological traceability.

For an original two-range measurement model, let the same stable electric current $I$ produceEliminating $I$ givesAn overlap experiment therefore needs enough distinct, stable input levels to distinguish relative gain from offset. One common input gives one equation for $a$ and $c$ and cannot identify both without additional information.

For example, suppose $G_L=0.10V/nA$, $b_L=0.01\mathrm{V}$, $G_H=0.50V/nA$ and $b_H=0.02\mathrm{V}$. Currents of $1$, $2$ and $3\mathrm{nA}$ give output pairs $(0.11,0.52)$, $(0.21,1.02)$ and $(0.31,1.52)\mathrm{V}$. They satisfy $V_H=5V_L-0.03\mathrm{V}$. Dividing raw outputs by their nominal gains without removing the offsets would not give consistent current estimates.

A practical test can drive a switchable transimpedance amplifier from a stable current source, record multiple overlap levels, and alternate the setting order to reveal instrumental drift. Avoid amplifier clipping, allow settling time, and retain the bandwidth and temperature conditions. The manufacturer tables show why range-dependent noise and gain drift must be included rather than assumed equal.^[ref]

At each level, compare corrected estimates $I_H$ and $I_L$ through their difference $d=I_H-I_L$. The measurement uncertainty of that difference depends on covariance:A common reference can move both estimates together and disappear from this difference. Thus good overlap agreement checks a relative relationship; it cannot by itself exclude a shared calibration error. The general uncertainty-combination law explicitly retains correlated contributions.^[ref]

Range-switching hysteresis uses different thresholds for changing from a more sensitive instrument range to a less sensitive one and for changing back. Within the intervening interval, the current range is retained. The selected range therefore depends on the previous state as well as the present measurement indication. This prevents small fluctuations near a single boundary from repeatedly triggering gain ranging.

A concrete electronic analogue is an electronic comparator with two switching thresholds. The primary circuit example specifies upper and lower thresholds of $2.5\mathrm{V}$ and $2.2\mathrm{V}$ and explains why one threshold can produce unwanted transitions with a noisy or slowly changing input. Those are comparator circuit values, not specifications for an autoranging measurement instrument.^[ref]

For an original two-range current measurement, let $H$ denote high sensitivity and $L$ low sensitivity. Use an input-current estimate $x$, expressed on the same calibrated scale in either range. Switch $H\to L$ when $x\ge 8\mathrm{nA}$, switch $L\to H$ when $x\le 6\mathrm{nA}$, and otherwise keep the previous state. The hysteresis interval has width $2\mathrm{nA}$.

Starting in $H$, the indication sequence $(7.9,8.1,7.9,8.1,5.9)\mathrm{nA}$ selects $(H,L,L,L,H)$. A single threshold at $8\mathrm{nA}$ would instead select $(H,L,H,L,H)$ and switch at every step after the first. At a steady underlying $7\mathrm{nA}$ with bounded readout fluctuations of $\pm 0.2\mathrm{nA}$, neither boundary is crossed; either previous range remains selected.

The thresholds should lie within the usable overlap established by range-overlap calibration. If the two ranges give inconsistent current estimates, a switch can itself move the indication across the opposite threshold and create a cycle. The comparison must therefore use the corrected measurand estimate, not identical raw output voltages from different gains.

Hysteresis is not temporal averaging and does not ensure that the new reading has settled. Reserve headroom for a rapidly rising input, reject readings during the required settling time, and verify that the less sensitive range still meets its noise-limited detection threshold. An input that jumps beyond the current range can cause amplifier clipping before the controller reacts, even when its two-threshold rule is correct.

The step response describes a sudden persistent input change. Suppose the input becomes a constant $V_s$ at $t=0$, and the initial capacitor voltage is $v_0$. Finite electric current leaves that voltage continuous. Subtract $V_s$ from the RC low-pass filter equation: the remaining error satisfies $\tau de/dt+e=0$. Separating variables givesFor an initially uncharged capacitor and a $0$ to $3\mathrm{V}$ step in the constructed example,At one time constant, $t=103.4\mu \mathrm{s}$, the voltage is $1.8964\mathrm{V}$, about $63.2\%$ of its final value. The voltage error then continues its exponential decay. This is different from a pure delay: the output starts changing immediately, yet takes time to approach the target.

Define settling time relative to the total voltage change: require $|v_{\mathrm{out}}-V_s|\le ϵ|v_0-V_s|$ thereafter, with $0<ϵ<1$. The exponential decay is monotone, soFor $1\%$ settling, $t_{0.01}=4.6052\tau \approx 476.17\mu \mathrm{s}$; the $10\%$ to $90\%$ rise takes $\tau \ln 9\approx 227.19\mu \mathrm{s}$. Reducing the cutoff frequency by doubling $C$ doubles both times. This reciprocal relation is the useful connection between signal bandwidth and response speed: stronger suppression of rapid noise comes with slower tracking of wanted changes.

Rise time measures how long a response takes to traverse a specified fraction of its transition from an initial level to a final level. A common convention is the 10–90-percent rise time. Normalize an upward transition bywhere $t_p$ is the crossing time at normalized level $p$. The percentages and crossing rule must be stated, especially when a waveform overshoots, rings or crosses a threshold repeatedly.^[ref]

For an original first-order low-pass filter with time constant $\tau$, its normalized response to a delayed ideal step is $z(t)=1-e^{-(t-t_d)/\tau }$ for $t\ge t_d$. Solving for the threshold givesA pure delay changes both crossings by the same amount and therefore does not change their difference. For $\tau =1\mathrm{ms}$, the crossings occur about 0.105 and 2.303 milliseconds after the response begins, giving a 2.197-millisecond rise time.

The same one-pole model has a minus-3-decibel bandwidth $B=1/(2\pi \tau )$, soThe familiar $t_r\approx 0.35/B$ relation therefore follows from this model and these thresholds. It is not a universal identity for every amplifier or oscilloscope. Different frequency responses can share a minus-3-decibel bandwidth while producing different transients.^[ref]

Rise time also differs from settling time. The example reaches 90 percent at $2.303\tau$ but does not reach the one-percent error band until $4.605\tau$. A waveform can rise quickly and then ring outside its required final tolerance.

To measure a device, apply a characterized step, record the response through the probe and oscilloscope, estimate the initial and final levels, and determine the stated threshold crossings. The source, device, probe and readout all contribute to the recorded waveform. Probe loading, limited sampling and noise can change the inferred crossings; a displayed fast edge is not automatically the device's undisturbed response.^[ref]

The sinusoidal steady state of a stable linear time-invariant system driven by a sinusoid is its persistent response at the driving frequency, after the relevant transient response has decayed. A phasor calculation finds that response; it does not generally satisfy the circuit's switch-on initial conditions. The late-time assumption is explicit in Berggren's treatment of sinusoidal circuit analysis.^[ref]

Using peak phasors, write the source as $v_s(t)=\mathrm{Re}({\hat{V}}_s{e}^{i\omega t})$. If $H(i\omega )$ is the relevant transfer function, its particular solution isThe complete response adds a homogeneous solution determined by the initial conditions. Approaching the displayed expression requires the modes affecting the measured output to decay; an unchanged amplitude and a fixed phase shift then characterize its frequency response.

For an original turn-on example, drive a series RC circuit with $v_s(t)=V_0\cos \omega t$ for $t\ge 0$, measure the capacitor's electric potential difference, and take $v_C(0)=0$. With time constant $\tau =RC>0$, Kirchhoff's voltage law gives the first-order linear differential equationPut $x=\omega \tau$. Substitution verifies the sinusoidal particular solution $V_0(\cos \omega t+x\sin \omega t)/(1+x^2)$. Adding $A{e}^{-t/\tau }$ and imposing the initial condition fixes $A=-V_0/(1+x^2)$, givingIts sinusoidal part has amplitude $V_0/\sqrt{1+x^2}$ and lags by $\arctan x$. The exponential settling bound for RC sinusoidal turn-on, relative to that amplitude, isWith chosen $R=1\mathrm{k\Omega }$, $C=1\mu \mathrm{F}$, $V_0=1\mathrm{V}$ and $\omega =1000rad/s$, we have $\tau =1\mathrm{ms}$ and $x=1$. At turn-on the phasor approximation predicts 0.5 volt while the capacitor is initially at zero. After 5 milliseconds, their difference is about 3.37 millivolts, or 0.476 percent of the sinusoidal amplitude. A 1 percent settling time threshold for this specific switch-on is about $4.259\tau$.

A useful measurement compares a long, low-frequency square-wave response with sinusoidal responses at several frequencies. An oscilloscope can identify the exponential decay and then measure amplitude ratios and phase shifts. MIT's laboratory uses that sequence to connect the time constant to the sinusoidal response; probe loading belongs in the measured circuit model.^[ref]

Undamped resonant forcing has no bounded sinusoidal steady state. For an ideal series LC circuit, let $q$ be electric charge and ${\omega }_0=1/\sqrt{LC}$. At resonance, the equationhas the particular solution $q_p(t)=V_{0}t\sin ({\omega }_{0}t)/(2L{\omega }_0)$. Its growing envelope prevents a bounded attracting sinusoidal response. This ideal counterexample explains why decay of the relevant homogeneous solutions is part of the physical steady-state assumption.

Unidirectional positioning repeatability describes the scatter of repeated returns to one commanded position when the final approach direction is held fixed. Specify the complete reset-and-return procedure so that each trial approaches under comparable conditions. PI distinguishes this directional scatter from its summaries that include reversal effects.^[ref]

For an original example, always approach from below. At a 2 mm target, suppose the position errors are $-0.4,-0.2,0.0\mu \mathrm{m}$; at an 8 mm target they are $1.6,1.8,2.0\mu \mathrm{m}$. The means are $-0.2$ and $1.8\mu \mathrm{m}$, respectively, while both sample standard deviations areThe same repeatability statistic accompanies quite different mean errors along the travel. A common shift of every reading at a target changes its mean but not its within-target spread.

For illustration, an explicitly declared specification taking the largest tested directional sample deviation would report $0.2\mu \mathrm{m}$ for these two data sets. A specification using a full width or a multiple of that deviation would have a different numerical value. PI's own published convention states which directional spreads and target aggregation it uses; compare like definitions rather than only their units.^[ref]

Observe final platform motion with an independent position readout, not just the motor's rotary encoder. Allow the chosen settling time and keep load and environment controlled. Consistently approaching from one side does not establish bidirectional positioning repeatability, and a small return scatter does not show that a small incremental command can be executed: that is the separate question of minimum incremental motion.