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The capacitive bridging complex-error bandwidth is the largest frequency over which a receiving port preserves the source phasor within a specified complex relative gain error. The criterion includes phase shift as well as amplitude. A large DC input impedance alone cannot establish this bandwidth when input capacitance matters. The manufacturer's input models explicitly include resistive and capacitive paths.^[ref]

For the scalar impedance bridging with capacitive input model, assume a positive real source electrical resistance $R_s$, receiver resistance $R_{\mathrm{in}}>0$, and input capacitance $C_{\mathrm{in}}>0$ in parallel. Let $V_s$ be the unloaded source phasor. A voltage divider givesFor tolerance $0<\eta <1$, the complex relative gain error isIts square is strictly increasing in $x^2$, since its derivative is $(1+2g)/[(1+g{)}^2+x^2{]}^2>0$. Thus satisfying $E(f)\le \eta$ defines a continuous frequency interval starting at DC, whenever it is feasible. Rearranging givesThe DC error floor is ${\eta }_0=g/(1+g)$. If $\eta <{\eta }_0$, no frequency meets the tolerance; if $\eta ={\eta }_0$, only DC does. The boxed positive bandwidth applies when $\eta >{\eta }_0$.

For an original near-limit example, choose $R_s=10\mathrm{k\Omega }$, $R_{\mathrm{in}}=1\mathrm{M\Omega }$ and $C_{\mathrm{in}}=100\mathrm{pF}$. Then $g=0.01$ and the DC error is about 0.9901 percent. A 1 percent full-complex tolerance leaves onlyAn amplitude-only 1 percent criterion, $|H|\ge 0.99$, instead allows about $2273.47\mathrm{Hz}$. The half-power (approximately $-3\mathrm{dB}$) signal bandwidth, relative to DC, is $(1+g)/(2\pi R_s{C}_{\mathrm{in}})\approx 160.75\mathrm{kHz}$. These are three distinct specifications of this same ideal model.

The square-root collapse of capacitive bridging bandwidth near the DC error floor follows by factoring the numerator:For fixed $g>0$ and $\eta \downarrow {\eta }_0$,This asymptotic equivalence explains the sharp loss of usable bandwidth when the resistive divider has already consumed nearly all the error allowance. Increasing $R_{\mathrm{in}}$ or reducing $R_s$ changes that floor; reducing only $C_{\mathrm{in}}$ rescales a feasible bandwidth but cannot remove infeasibility at DC.

The criterion also controls a complete waveform. Suppose a source waveform $v_s$ is both a square-integrable function and a bandlimited function, with spectrum supported where $|H-1|\le \eta$. For the corresponding zero-state response $v_L$, the Fourier transform of $v_L-v_s$ is $(H-1){\hat{v}}_s$. The Plancherel theorem, or the norm bound for a Fourier multiplier on L2, givesHere the Fourier transform uses the frequency-in-hertz convention, so the displayed Plancherel theorem has no extra normalization factor. This bound includes waveform changes caused by phase. It applies to the zero-state response of the stated frequency response. Any additional homogeneous solution from initially stored energy contributes its own error.

A bench check therefore needs both amplitude ratio and phase shift, with probe and cable capacitance included in the port model. Magnitude-only observations determine $|H|$ without determining $|1-H|$. The formula describes this chosen passive input network; additional amplifier dynamics require their own transfer function.

The crossover from square-root to linear capacitive bridging bandwidth scaling occurs when excess error allowance becomes comparable with the resistive error floor itself. In square-root collapse of capacitive bridging bandwidth near the DC error floor, put ${\eta }_0=g/(1+g)$, $g=R_s/R_{\mathrm{in}}$, $\Delta =\eta -{\eta }_0>0$, and ${\tau }_s=R_s{C}_{\mathrm{in}}$. For positive ${\eta }_0$, introduce the dimensionless ratio $\xi =\Delta /{\eta }_0$. The exact bandwidth can then be writtenThis single expression distinguishes the two regimes without exchanging small-parameter limits silently.

When $\Delta \ll {\eta }_0$ and the tolerance is away from one, the factor $\sqrt{\xi (2+\xi )}$ is approximately $\sqrt{2\xi }$: bandwidth scales as $\sqrt{\Delta }$. When ${\eta }_0\ll \Delta \ll 1$, that factor is approximately $\xi$, while the denominator is close to one: bandwidth is approximately $(1+g)\Delta /(2\pi {\tau }_s)$. The exactly zero-floor model has the corresponding small-tolerance linear behavior.

For an original dimensionless comparison, take ${\eta }_0=10^{-4}$. At $\xi =0.01$, replacing $\sqrt{\xi (2+\xi )}$ by $\sqrt{2\xi }$ changes the numerator by only about $0.25\%$. At $\xi =100$, replacing it by $\xi$ changes the numerator by about $1\%$; here the total tolerance is $0.0101$, so the denominator differs little from one. The tolerance is small in both cases, but a different approximation is useful because its size relative to the floor differs.

In component terms, increasing $R_{\mathrm{in}}$ moves the crossover by lowering ${\eta }_0$; changing only $C_{\mathrm{in}}$ rescales the frequency axis through ${\tau }_s$. A sweep of receiving electrical resistance and input capacitance can therefore separate a floor effect from a frequency-scale effect. The resistive/shunt-capacitive manufacturer model motivates the apparatus assumptions.^[ref] The specification remains a full complex relative gain error, including phase shift, rather than only an amplitude or half-power threshold.

An experimental condensed matter physicist who coauthored an early phase-sensitive test of the pairing state in a cuprate superconductor. Interference in devices joining the cuprate to lead tested a relative phase shift predicted by d-wave superconductivity. The original report discusses device asymmetry and trapped vortices, showing why several observations must be assessed together when inferring the pairing symmetry.^[ref]

Electronic gain expresses how an electronic circuit scales a specified input signal quantity into an output quantity. For a linearized voltage amplifier around its operating point, the voltage gain is $A_v=\delta v_{\mathrm{out}}/\delta v_{\mathrm{in}}$. At a sinusoidal frequency it can be a complex phasor ratio, specifying both amplitude ratio and phase shift. Power gain instead compares the average power delivered to the load with the average signal power entering the amplifier:State the input/output quantities, frequency, load and operating conditions when specifying gain. A transimpedance amplifier relates output electric potential difference to input current and has a dimensional transfer ratio in ohms.

For an original circuit example, use an inverting operational amplifier with input resistor $R_1=10\mathrm{k\Omega }$, feedback resistor $R_2=100\mathrm{k\Omega }$ and load $R_L=1\mathrm{k\Omega }$. In its ideal negative-feedback linear regime,An input of $50\mathrm{mV}$ root mean square therefore gives $0.50\mathrm{V}$ root mean square at the output, with a phase inversion. The manufacturer's primary explanation derives this resistance-ratio gain using the input and feedback currents.^[ref] Suitable supplies, output-current capability and amplifier output range are assumptions of this worked model, not specifications of a particular device.

Here the input resistance is $R_1$ in the ideal model. For resistive ports and root-mean-square signal amplitudes,The example input power is $0.25\mu \mathrm{W}$ and load power $0.25\mathrm{mW}$, giving $G_P=1000$. The additional output energy comes from the amplifier's supply, controlled by the input signal. The impedance condition for converting voltage gain to power gain is essential: voltage magnitude alone does not specify load power.

Voltage gain is conventionally expressed as $20{\log }_{10}|A_v|$ decibels, while power gain uses $10{\log }_{10}{G}_P$. This example has voltage gain $20\mathrm{dB}$ and power gain $30\mathrm{dB}$; the two expressions agree when the resistive input and load impedances are equal. A negative voltage gain indicates phase inversion; its magnitude enters the logarithm.

A direct test drives a small sine wave into the input and measures the actual terminal signals with an oscilloscope. Sweeping frequency reveals the frequency response; increasing amplitude reveals amplifier clipping or other departure from the assumed linear gain. Keep the load and measurement impedances specified. Gain can amplify input noise along with signal, so the signal-to-noise ratio requires its own measurement.

The impedance magnitude of a linear electrical circuit in sinusoidal steady state is the ratio of voltage amplitude to electric current amplitude. For complex impedance $Z=R+iX$, it is $|Z|=\sqrt{R^2+X^2}$, whereas the dissipative electrical resistance is $\mathrm{Re}Z=R$. Measuring only amplitudes does not separately determine $R$ and $X$; a phase shift measurement supplies the missing information.

Input capacitance describes the capacitive part of an electrical port's small-signal input impedance, under stated bias, frequency, and termination conditions. In a simple parallel model with the $e^{i\omega t}$ convention, its electrical admittance isThe electric current drawn by the capacitance grows with angular frequency even if the DC input resistance is enormous. Analog Devices distinguishes differential input paths from common-mode paths to ground: a quoted capacitance must identify which terminals are driven and how the remaining circuit is connected.^[ref]

For an original loading example, let $R_{\mathrm{in}}=10\mathrm{M\Omega }$ and $C_{\mathrm{in}}=20\mathrm{pF}$. At $10\mathrm{MHz}$, the capacitive reactance magnitude is only $1/(2\pi f{C}_{\mathrm{in}})\approx 796\Omega$. A source with electrical resistance $R_s=1\mathrm{k\Omega }$ therefore has loaded-to-unloaded phasor ratioThe DC ratio is about $0.99990$. Thus a resistance-only reading would miss a substantial high-frequency amplitude reduction. The numbers are chosen examples, not a specification for a named device.

For this same model, the source-to-input time constant is $(R_s\Vert R_{\mathrm{in}})C_{\mathrm{in}}$ and the DC-normalized half-power signal bandwidth is its reciprocal divided by $2\pi$. A complex-error requirement is instead governed by capacitive bridging complex-error bandwidth; these specifications answer different questions.

For input capacitance inferred from measured admittance, apply a small sinusoidal electric potential difference, measure the input electric current and its phase shift, and infer $\mathrm{Im}(I/V)/\omega$ within the parallel-capacitance approximation. Fixture, cable, and probe capacitances contribute to the measured port unless removed or included explicitly. Dependence on bias or frequency can make a single fitted $C_{\mathrm{in}}$ an inadequate model.

Lloyd's mirror produces interference between light arriving directly from a narrow source and light reflected by a plane mirror, which appears to originate at the source's virtual image. With source distance $h$ from the mirror, the effective source separation is $2h$. A reflection phase shift close to $\pi$ reverses the bright/dark condition relative to two equally phased sources. Spatial filtering, suitable temporal coherence, and direct/reflected beam overlap remain necessary.

An atomic physicist who coauthored a precision gravitational constant measurement using atom interferometry. The experiment required modeling the atomic trajectories and source masses to turn an observed phase shift into a value of the constant.^[ref]

Identity: www.unibo.it/sitoweb/marco.prevedelli

The large rises mark the opening of Ga and As K-shell photoionization channels at their respective thresholds. Above each edge, the outgoing photoelectron interferes with waves backscattered by neighboring atoms; that interference modulates the absorption transition probability. Near-edge resonances and multiple scattering require a different level of treatment from the simple high-energy single-scattering formula. Ga has four As nearest neighbors, while As has four Ga nearest neighbors, so the two edges sample different central atoms and scattering factors even though they probe the same nearest-neighbor bond lengths.

Write $k=\sqrt{2m_e(E-E_0)}/\hslash$ and $\chi =(\mu -{\mu }_0)/{\mu }_0$ relative to a smooth atomic background. The phase $2k{R}_j+{\varphi }_j(k)$ includes the outward-and-return propagation distance and atomic phase shifts. The factor $R_j^{-2}$ describes geometrical spreading; the backscattering amplitude depends on the neighbor species. The exponential $e^{-2R_j/\lambda (k)}$ accounts for loss of coherent photoelectron propagation. Averaging over Gaussian bond-length fluctuations gives $⟨e^{2ik\delta R}⟩=e^{-2k^2{\sigma }_j^2}$.

The printed polarization factor needs a notation correction: for K-shell dipole absorption it is proportional to $({\hat{\mathbf{R}}}_j\cdot \mathbf{ϵ}{)}^2={\cos }^2{\theta }_j$, not literally the cosine of a dimensional vector dot product. Four equivalent tetrahedral bonds have an isotropic sum of these projection weights. If the sum is over shells rather than individual neighbors, it also needs the coordination number. A common conventional amplitude contains $S_0^2{f}_j(k)/(k{R}_j^2)$; omitted normalization and $1/k$ factors must be absorbed in the printed $f_B$ convention rather than silently changing its dimensions.

These are the ingredients of extended X-ray absorption fine structure. A shell fit uses calculated or reference scattering amplitudes and phases, and refines bond distances, coordination and mean-square relative disorder together with an edge-energy correction. Fourier transformation separates approximate distance contributions, but the uncorrected peak position is shifted by the scattering phase and is not an exact bond length.^[ref]

Sample A has the larger Ga–As bond length. Its oscillations advance slightly faster with $k$: at higher $k$ its corresponding maxima lie to the left of B's. Reading several full cycles gives approximate periods $\Delta k_A\simeq 1.47{\mathring{\mathrm{A}}}^{-1}$ and $\Delta k_B\simeq 1.50{\mathring{\mathrm{A}}}^{-1}$. In the constant/common-phase-slope estimate, $R\propto 1/\Delta k$, soThis is a rough percent-level difference, not a precise fitted strain. Nonconstant atomic phase shifts affect the absolute distance inferred from a period and also limit this fractional estimate. The graph does not establish which alloy has which indium concentration or how much its bulk lattice parameter changed. Local Ga–As relaxation need not follow a uniform bulk strain exactly.

Low temperature reduces thermal relative motion, preserving high-$k$ oscillations and separating small phase changes more clearly. It does not remove zero-point motion or static alloy disorder. To obtain an illustrative disorder estimate, take the high-$k$ envelope of the plotted $k\chi (k)$ to fall from about $0.19$ at $k=7.5{\mathring{\mathrm{A}}}^{-1}$ to $0.12$ at $k=10.4{\mathring{\mathrm{A}}}^{-1}$. If other envelope factors vary slowly over this interval,The value is only an order-of-magnitude, model-dependent reading: the backscattering amplitude, mean free path and normalization have not been independently removed. In particular, ${\sigma }^2$ is the bond-projected mean-square relative displacement,including correlations and static disorder. It is not uniquely either atom's mean-square displacement. Equal uncorrelated projected displacements would each contribute ${\sigma }^2/2$, but that extra assumption is not supplied by the figure.

Take $u>0$ to mean that the mirror recedes along the incident plane electromagnetic wave. Put $\beta =u/c$, with $|\beta |<1$. The moving-mirror Doppler shift can be obtained directly by matching wave phase on the moving surface. Write the incident wave phase as $k_{i}z-{\omega }_{i}t$ and the reflected wave phase as $-k_{r}z-{\omega }_{r}t$, where $k_i={\omega }_i/c$ and $k_r={\omega }_r/c$ are positive wavenumbers. At the mirror, $z=z_0+ut$, the two phases must have the same time dependence; their constant offset incorporates the reflection phase shift. ThusConsequently the observed reflected angular frequency and ordinary frequency areThis is a redshift for a receding mirror and a blueshift for an approaching mirror. At small velocity, ${\omega }_r\simeq {\omega }_i(1-2u/c)$.

Equivalently, the relativistic Doppler effect gives ${\omega }_i^{\prime }=\gamma (1-\beta ){\omega }_i$ in the mirror rest inertial frame. Reflection there reverses propagation and preserves frequency. Transforming the backward-going reflected wave to the laboratory gives another factor $\gamma (1-\beta )$, so ${\omega }_r={\gamma }^2(1-\beta {)}^2{\omega }_i={\omega }_i(1-\beta )/(1+\beta )$. A single Doppler factor would miss the effect of the return path.

The output is measured across the capacitor. The voltage divider gives the RC low-pass filter transfer functionThereforeThe amplitude ratio tends to $1$ and the phase shift to zero as $\omega \to 0$. For $\omega \to \infty$, the amplitude ratio is asymptotic to $1/(\omega RC)$ and the phase shift tends to $-\pi /2$.

The half-power amplitude condition for the RC low-pass filter gives $1+(\omega RC{)}^2=2$. Since angular frequency is positive,At this cutoff angular frequency, the output amplitude is smaller by $1/\sqrt{2}$ and the phase shift is $-\pi /4$.

Use complex phasors with time dependence $e^{i\omega t}$ and assume an unloaded output. The capacitor has electrical impedance $Z_C=1/(i\omega C)$. The voltage divider formed with the resistor gives the transfer functionFor input $V_i(t)=V_{i0}\cos \omega t$, the output amplitude and phase shift areso $V_o(t)=|{\tilde{V}}_o|\cos (\omega t-{\tan }^{-1}(\omega RC))$.

Here $\tau =RC=16.0\mu \mathrm{s}$, and the cutoff frequency is $f_c=1/(2\pi RC)=9.947\mathrm{kHz}$. This is an RC low-pass filter: at small frequency the capacitor has large electrical impedance and the output nearly equals the input; at large frequency it increasingly shunts the signal to ground. The amplitude then falls as $1/f$, and the phase shift approaches $-90^{\circ }$.

No numerical input amplitude is supplied, so the output voltages are specified as fractions of $V_{i0}$. An application is suppression of high-frequency electrical noise on a slowly varying signal. A finite load changes the transfer function and must be included if its electrical impedance is comparable with the circuit's output electrical impedance.

For an ideal operational amplifier under negative feedback, the grounded non-inverting terminal makes the inverting terminal a virtual ground, and no electric current enters either input. The feedback resistor and capacitor have parallel electrical impedanceKirchhoff's current law at the inverting node gives ${\tilde{V}}_i/R_1+{\tilde{V}}_o/Z_f=0$, and henceWith the stated components, $R_{2}C=16.0\mu \mathrm{s}$ and $R_2/R_1=5$. Thus this inverting low-pass amplifier has exactly the same ideal cutoff frequency and normalized frequency response as the passive RC low-pass filter, with output multiplied by $-5$:Its amplitude ratios at the three stated frequencies are $4.99975$, $3.52616$, and $0.97560$. One principal choice of phase shift is $180^{\circ }-{\tan }^{-1}(\omega \tau )$: respectively $179.424^{\circ }$, $134.848^{\circ }$, and $101.252^{\circ }$. Adding or subtracting $360^{\circ }$ describes the same output; the extra inversion must be retained.

A mercury-vapor lamp is an extended, spontaneously emitting source, so practical interference requires spatial filtering and spectral selection. A narrow illuminated source slit gives sufficient spatial coherence; selecting one strong spectral line improves temporal coherence and avoids overlapping differently spaced interference fringes.

In the classical double-slit experiment, both apertures are illuminated by the same narrow source. The arrangement is symmetric, permits direct measurement and adjustment of slit separation, and has equal optical materials in the two paths. Its main drawbacks are low transmitted intensity through narrow apertures, the need for a well-made closely spaced slit pair and alignment, and the finite single-slit diffraction envelope. A broad source washes out interference fringes.

Lloyd's mirror uses the direct wave from a narrow source and the reflected wave, which appears to come from its virtual image behind a plane mirror. It requires only one physical source slit; the mirror can be adjusted to make a small effective source separation without manufacturing a double aperture. Only the direct-reflected overlap region shows interference fringes. Finite mirror size, alignment, surface quality, unequal intensities and the polarization-dependent reflection coefficient can reduce visibility. For the usual grazing reflection from glass, the reflected electric wave acquires approximately a $\pi$ phase shift, so the zero path-difference position is dark rather than the bright central fringe of the symmetric classical double-slit experiment. Both arrangements still require adequate spatial coherence and temporal coherence.

The voltage divider relation for the capacitor output gives the transfer functionConsequently the relative wave amplitude isFor input $V_0\cos \omega t$, the output is $V_0|H|\cos (\omega t+\varphi )$, where the continuous phase shift isFor positive frequency and $R>0$, this phase shift goes from $0$ towards $-\pi$, passing through $-\pi /2$ at ${\omega }_0$. A one-argument arctangent without a quadrant correction would wrongly predict a positive phase shift above ${\omega }_0$. The capacitor follows a slowly varying input voltage, while its output falls as $1/({\omega }^{2}LC)$ at high frequency.

The single secondary turn links the primary magnetic flux ${\mu }_{0}n\pi a^2{I}_p$, so its mutual inductance with the primary isChoose positive winding directions so that positive secondary electric current would reinforce the primary magnetic flux. Faraday's law and Kirchhoff's voltage law then giveAfter the transient proportional to $e^{-Rt/L_s}$ has decayed, put $\alpha ={\tan }^{-1}(\omega L_s/R)$. The steady electric current isIts wave amplitude is $\omega M{I}_0/\sqrt{R^2+(\omega L_s{)}^2}$ and it lags the induced electromotive force by $\alpha$. Relative to the primary $I_0\cos \omega t$, its phase shift is $-\pi /2-\alpha$, modulo $2\pi$. Reversing the chosen secondary winding direction adds $\pi$ to this signed phase shift and leaves all power predictions unchanged.

The lamp's mean Joule heating isFor $\omega L_s\ll R$ the response is nearly in phase with the induced electromotive force; for $\omega L_s\gg R$ the secondary electric current nearly opposes the primary electric current in the stated winding convention.

A bubble is a thin water layer surrounded by air. At a locally flat patch, let its thickness be $d$, its refractive index be $n$, and the internal angle of refraction be ${\theta }_t$. The two principal reflected light contributions are the reflection at the first surface and the contribution that enters the layer, reflects at its second surface and leaves again. The first reflection, from lower to higher refractive index, introduces a $\pi$ phase shift; the second does not. Their optical path difference is $2nd\cos {\theta }_t$, so their phase difference is, modulo $2\pi$,where ${\lambda }_0$ is the vacuum wavelength. Constructive interference in reflected light therefore occurs approximately whenDestructive interference occurs at $2nd\cos {\theta }_t=m{\lambda }_0$. In the weak-reflection approximation, the reflected light intensity at each wavelength is proportional to ${\sin }^2(2\pi nd\cos {\theta }_t/{\lambda }_0)$. This is a smooth spectral filter, with broad maxima and minima: it does not select three pure wavelengths corresponding to display primaries. Ordinary illumination contains many visible wavelengths, and the eye combines their remaining contributions through color vision. The result is a mixture of colours, often pastel or desaturated.

The layer thickness varies over the bubble because of drainage and flow; the bubble's curvature also changes the local angle of refraction and the viewed optical path difference. The spectral filter therefore changes continuously across the surface and in time. Very thin regions become dark because the two reflected contributions nearly cancel across the visible optical spectrum. Thin-film interference explains both the changing colours and this dark limit.

The ray sketch shows a local patch and is not to scale. The plotted curves use a chosen constant $n=1.33$ at normal incidence and illustrative thicknesses; they show the spectral filter rather than calibrated perceived colours.

Normalize the central slit amplitude to one. The two outer amplitudes are $1/2$ each, not $1/\sqrt{2}$, because the specified attenuation concerns amplitude. For centre spacing $d=5\lambda$, adjacent slits have phase difference $\delta =(2\pi d/\lambda )\sin \theta =10\pi \sin \theta$. The far-field amplitude and intensity are thereforewhere $I_0$ is the central slit's intensity alone in the same narrow-slit approximation. Principal maxima have $\sin \theta =m/5$ and height $4I_0$; between them, zeros occur at $\sin \theta =(2m+1)/10$. Only values in $[-1,1]$ are physical. The endpoints are grazing-direction limits of this ideal angular model.

The amplitude weights form a binomial pattern. They remove the secondary maxima that an equal-amplitude three-slit array would have between its principal peaks. No phase shift is needed: changing the coherent weights already changes the interference structure.

A neutral superfluid has a complex order parameter whose phase describes coherent particle transport. For a Bose condensate it is represented by a field amplitude; for a paired Fermi superfluid it is a pair amplitude. In a symmetry-breaking description, selecting its phase breaks the global particle-number $U(1)$ symmetry. A common phase shift leaves the energy unchanged, while a phase gradient costs energy and produces superfluid velocity. Long-wavelength phase oscillations give a gapless Goldstone mode, generally a sound wave when density fluctuations provide the conjugate dynamics. A finite isolated state with exact particle number need not have a nonzero field expectation: the phase-selected language describes the ordered limit and relative-phase coherence.

In a superconductor, the paired order parameter carries charge of magnitude $2e$ and couples to the electromagnetic field. The meaningful spatial phase combination is $\hslash \nabla \theta -q\mathbf{A}$, with signed pair charge $q$. Gauge symmetry is a redundancy, so “breaking local gauge symmetry” is not a literal breaking of a measurable physical symmetry. The gauge-invariant electromagnetic response includes the Meissner effect and a finite London penetration depth. Long-range Coulomb forces move the would-be neutral phase mode to a plasma scale through the Anderson-Higgs mechanism in a superconductor.

Additional symmetries can be broken by particular pairing states. Conventional isotropic spin-singlet pairing need not break spatial rotations or time reversal. A d-wave superconductor has a gap transforming nontrivially under crystal symmetries, and some triplet or multicomponent states can also break spin-rotation or time-reversal symmetries. These additional choices are not required by superconductivity itself. A nonzero pairing amplitude, phase coherence and electromagnetic screening describe related but distinct aspects of the ordered state.